Radio frequency transmitter having translational loop phase equalization

ABSTRACT

A Radio Frequency RF transmitter includes a translational loop architecture that supports non-constant envelope modulation types and includes by adjusting the envelope of the translational loop at the translational loop output. The RF transmitter includes a phase equalizer, an Intermediate Frequency (IF) modulator, a translational loop, an envelope time delay adjust block, an envelope adjust block, and a time delay calibration block. The phase equalizer receives a modulated baseband signal and phase equalizes the modulated baseband signal to produce a phase equalized modulated baseband signal. The IF modulator receives the phase equalized modulated baseband signal and produces a modulated IF signal having a non-constant envelope. The translational loop receives the modulated IF signal and produces a modulated RF signal having a constant envelope. The envelope time delay adjust block receives an envelope signal corresponding to the original modulated signal and produces a time delayed envelope signal based upon a time delay control signal. The envelope adjust block adjusts the modulated RF signal based upon the time delayed envelope signal to produce an envelope adjusted modulated RF signal. Finally, the time delay calibration block receives the envelope adjusted modulated RF signal and produces the time delay control signal.

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation in part to U.S. Ser. No. 10/066,843filed Feb. 4, 2002, which claims priority to U.S. ProvisionalApplication Ser. No 60/338,000, filed Dec. 7, 2001, the disclosure ofboth incorporated herein by reference.

FIELD OF THE INVENTION

This invention relates generally to wireless communications; and moreparticularly to the operation of a Radio Frequency (RF) transmitterwithin a component of a wireless communication system.

BACKGROUND OF THE INVENTION

The structure and operation of wireless communication systems isgenerally known. Examples of such wireless communication systems includecellular systems and wireless local area networks, among others.Equipment that is deployed in these communication systems is typicallybuilt to support standardized operations, i.e. operating standards.These operating standards prescribe particular carrier frequencies,channels, modulation types, baud rates, physical layer frame structures,MAC layer operations, link layer operations, etc. By complying with tothese operating standards, equipment interoperability is achieved.

In a cellular system, a governmental body licenses a frequency spectrumfor a corresponding geographic area (service area) that is used by alicensed system operator to provide wireless service within the servicearea. Based upon the licensed spectrum and the operating standardsemployed for the service area, the system operator deploys a pluralityof carrier frequencies (channels) within the frequency spectrum thatsupport the subscribers' subscriber units within the service area. Thesechannels are typically equally spaced across the licensed spectrum. Theseparation between adjacent carriers is defined by the operatingstandards and is selected to maximize the capacity supported within thelicensed spectrum without excessive interference. In most cases, severelimitations are placed upon the amount of adjacent channel interferencethat may be caused by transmissions on a particular channel.

In cellular systems, a plurality of base stations is distributed acrossthe service area. Each base station services wireless communicationswithin a respective cell. Each cell may be further subdivided into aplurality of sectors. In many cellular systems, e.g., GSM cellularsystems, each base station supports forward link communications (fromthe base station to subscriber units) on a first set of carrierfrequencies and reverse link communications (from subscriber units tothe base station) on a second set carrier frequencies. The first set andsecond set of carrier frequencies supported by the base station are asubset of all of the carrier frequencies within the licensed frequencyspectrum. In most, if not all cellular systems, carrier frequencies arereused so that interference between base stations using the same carrierfrequencies is minimized but so that system capacity is increased.Typically, base stations using the same carrier frequencies aregeographically separated so that minimal interference results.

Both base stations and subscriber units include Radio Frequency (RF)transmitters and RF receivers, together “RF transceivers.” RFtransceivers service the wireless links between the base stations andsubscriber units. The RF transmitter receives a baseband signal from abaseband processor, converts the baseband signal to an RF signal, andcouples the RF signal to an antenna for transmission. In most RFtransmitters, because of well-known limitations, the baseband signal isfirst converted to an Intermediate Frequency (IF) signal and then the IFsignal is converted to the RF signal. The RF receiver receives an RFsignal, converts the RF signal to an IF signal, and then converts the IFsignal to a baseband signal, which it then provides to the basebandprocessor.

The fast growth of the mobile communications market demands multi-bandRF transceivers that are small in size, low in cost, and have low powerconsumption. These market demands require that the architecture of theRF transceiver to be suitable for a high level of system integration ona single chip for reduced cost and miniaturized mobile device size. Lowpower consumption is very critical for increasing mobile device batterylife and is very important for small mobile devices that include smallbatteries. To meet these design challenges, some RF transmitters now usetranslational loop architecture to convert the IF signal to an RFsignal. Translational loop architectures are useful for constantenvelope modulated wireless systems, such as the new generation GlobalStandards for Mobile Communications (GSM) and General Packet RadioSystem (GPRS) phones that employ Gaussian Minimum Shift Keying (GMSK)modulation. However, so far, the translational loop architecture has notbeen successfully applied in systems that employ a non-constant envelopemodulation format, such as QPSK for CDMA (IS-95) and US-TDMA (IS-136)standardized systems, for 8-PSK for EDGE standard based mobile devices,and for mobile devices that support other non-constant envelopemodulation formats, such as 16 QAM, 32 QAM, 64 QAM, 128 QAM, etc.

Thus, there is a need in the art for a lower power consumption RFtransmitter that supports both constant envelope modulation formats andnon-constant envelope formats, among other shortcomings of the priordevices.

SUMMARY OF THE INVENTION

Thus, in order to overcome the above-described shortcomings as well asother shortcomings of the present devices and methodologies, an RFtransmitter constructed according to the present invention includes atranslational loop architecture that supports non-constant envelopemodulation types, e.g., QPSK, 8-PSK, 16 QAM, 32 QAM, 64 QAM, 128 QAM,etc. The translational loop architecture of the present inventionadjusts the envelope of the translational loop so that it supportsnon-constant envelope modulation types. The RF transmitter may becontained in a mobile device or a stationary device.

In particular, the RF transmitter includes an Intermediate Frequency(IF) modulator, a translational loop, an envelope time delay adjustblock, an envelope adjust block, and a time delay calibration block. TheIF modulator receives a modulated baseband signal and produces amodulated IF signal having a non-constant envelope. The translationalloop receives the modulated IF signal and produces a modulated RF signalhaving a constant envelope. The envelope time delay adjust blockreceives an envelope signal corresponding to the modulated signal andproduces a time delayed envelope signal based upon a time delay controlsignal. The envelope adjust block adjusts the modulated RF signal basedupon the time delayed envelope signal to produce an envelope adjustedmodulated RF signal. Finally, the time delay calibration block receivesthe envelope adjusted modulated RF signal and produces the time delaycontrol signal.

In one embodiment, the time delay calibration block includes a downconverter, an Analog to Digital Converter (ADC), a Low Pass Filter(LPF), a Band Pass Filter (BPF), and a level detector and control block.The down converter converts the envelope adjusted modulated RF signal toa complex baseband signal. The ADC samples the complex baseband signal.The LPF couples to the ADC and filters the complex baseband signal toproduce a LPF output. The BPF also couples to the ADC and filters thecomplex baseband signal to produce a BPF output. With this structure,the level detector and control block receives the LPF output and the BPFoutput and produces the time delay control signal based upon the LPFoutput and the BPF output. These components of the RF transmitter may beembodied within resources resident in a coupled baseband processor.

The time delay calibration block of the RF transmitter determines adesired channel power corresponding to the RF signal. The time delaycalibration block also determines an alternate channel power (oradjacent channel power) that is emitted by the translational loop thatcorresponds to an alternate channel (or adjacent channel). The timedelay calibration block then determines the time delay control signalbased upon a ratio of the channel power and the alternate channel power.With the time delay set properly, the envelope of the RF signal producedby the translational loop is substantially or fully matched with thephase of the RF signal produced by the translational loop. Resultantly,the alternate channel power (and the adjacent channel power) produced bythe translational loop is reduced/minimized to meet systematicrequirements.

The RF transmitter may also include an envelope detection block thatproduces the envelope signal. In one embodiment, the envelope detectionblock determines the envelope signal based upon the complex basebandsignal received from the baseband processor (not the complex basebandsignal generated within the time delay calibration block). In anotherembodiment, the envelope detection block determines the envelope signalbased upon the modulated IF signal. In still another embodiment, theenvelope detection block receives the envelope signal from a coupledbaseband processor.

In at least some embodiments, the envelope signal produced by theenvelope detection block is a digital signal while the time delayedenvelope signal is an analog signal. With this signal format, the timedelay block delays a digital envelope signal by a delay that is basedupon the time delay control signal. Then, a digital to analog converterreceives the output of the time delay block and produces the timedelayed envelope signal.

The translational loop may have a phase response that varies withfrequency such that the phase response of the translational loop isconstant within the desired channel but not constant within an adjacentchannel and/or in an alternate channel, i.e., the phase response is afunction of frequency within the adjacent channel and/or the alternatechannel. Because of this non-constant phase response of thetranslational loop, even if the envelope and phase at the output of thetranslational loop are perfectly matched for the desired channel, theywill be mismatched in the adjacent channel and/or alternate channel.Resultantly, the adjacent channel power and the alternate channel poweroutput by the translational loop may violate systematic requirements.

Thus, with another RF transmitter constructed according to the presentinvention, an incoming baseband signal (or IF signal) is phase equalizedby a phase equalizer to compensate for the non-constant phase responseof the translational loop. With this phase equalization performed priorto the translational loop, the non-constant phase response of thetranslational loop is overcome so that the adjacent channel power andalternate channel power output by the translational loop is furtherreduced.

Other features and advantages of the present invention will becomeapparent from the following detailed description of the invention madewith reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects and advantages of the presentinvention will be more fully understood when considered with respect tothe following detailed description, appended claims and accompanyingdrawings wherein:

FIG. 1A is a system diagram illustrating a cellular system within whichthe present invention is deployed;

FIG. 1B is a block diagram generally illustrating the structure of awireless device constructed according to the present invention;

FIG. 2 is a block diagram illustrating a Radio Frequency (RF)transmitter constructed according to the present invention;

FIG. 3 is a block diagram illustrating in more detail the RF transmitterof FIG. 2;

FIG. 4 is a block diagram illustrating another embodiment of the timedelay calibration block of FIG. 2 that performs the Alternate ChannelPower Ratio (ACPR) measurement operations;

FIG. 5A is a block diagram illustrating a low pass filter employed in anACPR measurement block constructed according to the present invention;

FIG. 5B is a block diagram illustrating a clock generation circuitconstructed according to the present invention;

FIG. 6 is a block diagram illustrating a power level detectorconstructed according to the present invention; and

FIG. 7 is a block diagram illustrating a time delay block of theenvelope time delay adjust block constructed according to the presentinvention;

FIG. 8 is a block diagram illustrating a subscriber unit constructedaccording to the present invention;

FIG. 9 is a logic diagram illustrating a method of operation accordingto the present invention;

FIG. 10A is a graph illustrating the power spectral density of an RFsignal generated by the RF transmitter of the present invention with anenvelope adjust time delay mismatch;

FIG. 10B is a graph illustrating the typical power spectral density ofan RF signal generated by the RF transmitter of the present inventionwith the envelope adjust time delay correctly set;

FIG. 11 is a graph illustrating the phase response of a translationalloop employed by an RF transmitter constructed according to the presentinvention;

FIG. 12 is a block diagram illustrating another embodiment of an RFtransmitter constructed according to the present invention;

FIG. 13 is a block diagram illustrating one embodiment of a phaseequalizer of the RF transmitter of FIG. 12 constructed according to thepresent invention;

FIG. 14 is a block diagram illustrating another embodiment of a phaseequalizer of the RF transmitter of FIG. 12 constructed according to thepresent invention;

FIG. 15 is a logic diagram illustrating a method for calibrating a phaseequalizer according to the present invention; and

FIG. 16 is a logic diagram illustrating operation of a phase equalizeraccording to the present invention.

DETAILED DESCRIPTION OF THE DRAWINGS

FIG. 1A is a system diagram illustrating a cellular system within whichthe present invention is deployed. The cellular system includes aplurality of base stations 102, 104, 106, 108, 110, and 112 that servicewireless communications within respective cells/sectors. The cellularsystem services wireless communications for a plurality of wirelesssubscriber units. These wireless subscriber units include wirelesshandsets 114, 118, 120, and 126, mobile computers 124 and 128, anddesktop computers 116 and 122. When wirelessly communicating, each ofthese subscriber units communicates with one (or more during handoff) ofthe base stations 102 through 112. Each of the subscriber units 114–128and each of the base station 102–112 include radio frequency (RF)circuitry constructed according to the present invention.

The RF circuitry of the present invention may be contained in any of thesubscriber units 114–128, any of the base stations 102–112 or in anotherwireless device, whether operating in a cellular system or not. Thus,for example, the teachings of the present invention may be applied towireless local area networks, two-way radios, satellite communicationdevices, or other devices that support wireless communications.

The RF circuitry of the present invention supports both constantenvelope and non-constant envelope modulation types. The RF transmittersection of this RF circuitry supports non-constant envelope modulationformats such as QPSK for CDMA (IS-95) and US-TDMA (IS-136) standardizedsystems and 8-PSK for EDGE standardized systems. The RF transmitter ofthe present invention also supports other non-constant envelopemodulation types, e.g., QPSK, 8-PSK, 16 QAM, 32 QAM, 64 QAM, 128 QAM,etc, whether standardized or not. The structure and operation of the RFtransmitter is described further with reference to FIGS. 2–10B

The structure and operation of the RF transmitter of the presentinvention may also be implemented to service other wirelesscommunications as well. For example, the RF transmitter may be used toservice premises based Wireless Local Area Network (WLAN)communications, e.g., IEEE 802.11a and IEEE 802.11b communications, andad-hoc peer-to-peer communications, e.g., Bluetooth. In a WLAN system,the structure would be similar to the structure shown in FIG. 1A but,instead of the base stations 102–112, the WLAN system would include aplurality of Wireless Access Points (WAPs). Each of these WAPs wouldservice a corresponding area within the serviced premises and wouldwirelessly communicate with serviced wireless devices. For peer-to-peercommunications, such as those serviced in Bluetooth applications, the RFtransmitter of the present invention would support communicationsbetween peer devices, e.g., lap top computer 124 and hand-held wirelessdevice 126.

FIG. 1B is a block diagram generally illustrating the structure of awireless device 150 constructed according to the present invention. Thegeneral structure of the wireless device 150 will be present in any ofthe wireless devices 114–128 illustrated in FIG. 1A. The wireless device150 includes a plurality of host device components 152 that service allrequirements of the wireless device 150 except for the RF requirementsof the wireless device 150. Of course, operations relating to the RFcommunications of the wireless device 150 will be partially performed bythe host device components 152.

Coupled to the host device components 152 is the RF interface 154. TheRF interface 154 services the RF communications of the host device 150and includes a baseband processor 160, an RF transmitter 156, and an RFreceiver 158. The RF transmitter 156 and the RF receiver 158 both coupleto an antenna 160. One particular structure of a host device isdescribed with reference to FIG. 8. Further, the teachings of thepresent invention are embodied within the RF transmitter 156 of the RFinterface 154.

FIG. 2 is a block diagram illustrating an RF transmitter 200 constructedaccording to the present invention. The RF transmitter 200 includes anI-Q Intermediate Frequency (IF) modulator 202 that receives an I-Qbaseband signal from a baseband processor and converts the I-Qrepresented baseband signal into a modulated IF signal. The modulated IFsignal produced by the I-Q IF modulator serves as the input to atranslational loop 204. The translational loop 204 produces an RF signalat the Transmit (TX) frequency that is received by an envelope adjustblock 206. Note that the modulated IF signal received by thetranslational loop 204 is a non-constant envelope signal (correspondingto a non-constant envelope modulation type) while the RF signal outputby the translational loop 204 is a constant envelope signal. Theenvelope adjust block 206 adjusts the envelope of the RF signal producedby the translational loop 204 based upon a time delayed envelope signal.The envelope adjust block 206 produces an envelope adjusted RF signalthat is amplified by Power Amplifier (PA) 208. The output of the PA 208is applied to an RF switch, a RF filter or a duplexer that couples theamplified envelope adjusted RF signal to an antenna.

An envelope detection block 210, an envelope time delay adjust block212, and a time delay calibration block 214 work in cooperation toproduce the time delayed envelope signal. Envelope detection block 210detects the envelope of the IF signal using one of three (or more)techniques/embodiments. In a first embodiment, the envelope detectionblock 210 receives an envelope indication from a coupled basebandprocessor. In a second embodiment, the envelope detection block 210measures I and Q components of the baseband signal at the input of theI-Q IF modulator to calculate the envelope. In a third embodiment, theenvelope detection block 210 measures the envelope of the IF signal atthe output of the I-Q IF modulator 202.

The envelope signal, which represents a magnitude of the modulatedsignal, is output by the envelope detection block 210 and received bythe envelope time delay adjust block 212. The envelope time delay adjustblock 212 outputs a time delayed envelope signal. The time delayedenvelope signal, with respect to the envelope signal, has been delayedso that it correctly corresponds to the RF signal output of thetranslational loop 204. Thus, the delay introduced by the envelopeadjust time delay 212 corresponds to the delay introduced by thetranslational loop 204, and in addition, the delay introduced by the I-QIF modulator 202 when the envelope is detected from the baseband signalat the input to the I-Q IF modulator 202. Thus, the RF signal output ofthe envelope adjust block 206 is a phase and magnitude matched RF signalhaving a non-constant envelope.

A time delay calibration block 214 adjusts the delay introduced into thedetected envelope by the envelope time delay adjust block 212. Theoperation of the time delay calibration block 214 will be described indetail with reference to FIGS. 3–6 and 9–10B. Generally speaking, thetime delay calibration block 214 adjusts the time delay introduced bythe envelope time delay adjust block 212 until the ratio of the signallevel within a subject channel to the signal level of an adjacentchannel (or an alternate channel) satisfies the operating requirementsof the RF transmitter 200.

FIG. 3 is a block diagram illustrating in more detail the RF transmitter200 of FIG. 2. As shown, the I-Q IF modulator 202 receives an IFreference signal. In the described embodiment, the IF reference signalis equal to ⅙ of the transmit frequency (TX) reference signal. A phaseshifter 304 receives the IF reference signal (I component) and producesa corresponding Q component by shifting in phase the I component by 90degrees. An I-Q mixer 302 separately mixes the I and Q components of thebaseband signal with the I and Q components of the IF reference signal.The output of the I-Q mixer 302 is summed and band pass filtered byblock 306, which produces the modulated IF signal.

The translational loop 204 includes a Phase and Frequency Detector (PFD)308 that receives the modulated IF signal and produces an output that isreceived by a charge pump 310. Low Pass Filter (LPF) 312 filters theoutput of the charge pump 310 and the output of the LPF 312 drives a TXVoltage Controlled Oscillator (VCO) 314. The output of the TX VCO 314serves as one input to the envelope adjust block 206, which is a mixerin the illustrated embodiment. The translational loop 204 also includesa mixer 316 that receives the output of the TX VCO 314 and a 7/6 TXreference signal from a local oscillator. The output of mixer 316 with ⅙of the TX frequency serves as a second input to the PFD 308.

The envelope detection block 210 implements the second embodimentdescribed above by calculating the I and Q components in a digitalformat. The envelope detection block 210 also includes processingoperations to determine the magnitude of the envelope by calculating thesquare root of (I²+Q²). When the envelope detection block 210 implementsthe third embodiment described above, it digitizes the modulated IFsignal produced at the output of the I-Q IF modulator and determines themagnitude thereof.

The envelope time delay adjust block 212 includes a time delay block 322and a Digital to Analog converter (DAC) 324. The time delay calibrationblock 214 includes a mixer 326 that mixes the RF signal produced bymixer 206 (envelope adjust block) with a local oscillator signal at afrequency equal to the TX carrier frequency to down convert themodulated TX signal to a complex baseband signal at the output of themixer 326. An Analog to Digital Converter (ADC) 328 receives the outputof the mixer 326 and converts the signal to a digital equivalent of thecomplex baseband signal. Both a Low Pass Filter (LPF) 330 and a BandPass Filter (BPF) 332 filter the complex baseband signal. As an exampleof an operation that would be performed in an EDGE standard basedcellular system, the LPF 330 would have a cutoff frequency (F_(c)) of200 kHz, and the BPF 332 would have an F_(o) of 500 kHz and a Bandwidthof 200 kHz.

The outputs of the LPF 330 and the BPF 332 are received by a leveldetermination and control block 334 that produces the time delay controlsignal that is an input to the time delay block 322. In the describedembodiment, the LPF 330 produces a signal that corresponds to a subjectchannel and the BPF 332 produces a signal that corresponds to analternate channel (or adjacent channel). The level determination andcontrol block 334 uses these signals to calculate an Alternate ChannelPower Ratio (ACPR) that is the ratio of the Channel Power (CP) to theAlternate Channel Power (ACP). The ACPR is then compared to an allowablelimit. If the ACPR meets this limit, the input to the time delay block322 is properly set. However, if the ACPR does not meet this limit, thelevel determination and control block 334 adjusts the time delay controlsignal to the time delay block 322 until the limit is met.

FIG. 4 is a block diagram illustrating another embodiment of the timedelay calibration block 214 of FIG. 2 that performs the ACPR measurementoperations. As shown in FIG. 4, for hardware efficient filtering of thealternate channel signal, the BPF 332 includes a DDFS Look Up Table(LUT) 402, a complex mixer 404, and a LPF 406. Further, the leveldetection and control block 334 includes a power level detector 410 anda time delay control signal block 412. The power level detector 410outputs detected levels of the CP and the ACP. Based upon a ratio ofthese measurements, the ACPR, the time delay control signal block 412produces the time delay control signal.

FIG. 5A is a block diagram illustrating a LPF employed in an ACPRmeasurement block constructed according to the present invention. TheLPF includes filter operation blocks 502, 506, 510, and 512 and gainoperator 504. The low pass filter of FIG. 5A may be employed forboth/either of LPFs 330 and 406 of FIG. 4.

FIG. 5B is a block diagram illustrating a clock generation circuitconstructed according to the present invention. As was described withreference to FIG. 3, various reference signals must be produced for theRF transmitter 200. These frequencies include 1/6 TX, TX, and 7/6 TX. Inorder to produce these frequencies, the clock generation circuit of FIG.5B includes a Local Oscillator 552 that produces a clock referencesignal. A Phase Locked Loop (PLL) 554 receives the clock referencesignal and produces a frequency-multiplied output based thereupon. Afirst divider 556 receives the frequency-multiplied output and dividesthe signal to produce the 7/6 TX reference signal. A second divider 558receives the frequency-multiplied output and divides the signal toproduce the 1/6 TX (IF) reference signal. Further, in order to producethe TX reference signal (for use by the time delay calibration block214), the 7/6 TX reference signal and the 1/6 TX reference signal aremixed via mixer 560 to produce the TX reference signal.

FIG. 6 is a block diagram illustrating a power level detectorconstructed according to the present invention. As is shown, the powerlevel detector includes a first mixer 602 that is employed to producethe Power Spectral Density (PSD) of the signal output by the LPF 330.Further, a second mixer 604 is employed to produce the PSD of the signaloutput by the BPF 332. Filtering blocks 606 and 610 filter the outputsof the mixers 602 and 604, respectively. Subsequently, the outputs offiltering blocks 606 and 610 are down sampled by down sampling blocks608 and 612, respectively, to produce the CP and ACP signals,respectively. Note that the input to the power level detector 410 is ata sample rate of 26 MHz and that the output of the power level detector410 is at a sample rate of 10 kHz.

FIG. 7 is a block diagram illustrating a time delay block 322 of theenvelope time delay adjust block 212 constructed according to thepresent invention. The time delay block 322 receives as its input theenvelope signal produced by the envelope detection block 210 and thetime delay control signal produced by the time delay calibration block214. The time delay block 322 includes a plurality of serially coupleddelay elements 702–712. Multiplexer 714 receives as its input theenvelope signal and the output of each of the serially coupled delayelements 702–712. The multiplexer 714 receives as its control input thetime delay control signal from the time delay calibration block 214. Asits output, the multiplexer 714 produces the time delayed envelopesignal. The delay introduced by the delay elements 702–712 is determinedbased upon expected minimum and maximum delays that are required tocompensate for the delay introduced by the translational loop 204 (andthe I-Q IF modulator 202).

FIG. 8 is a block diagram illustrating a subscriber unit 802 constructedaccording to the present invention. The subscriber unit 802 operateswithin a cellular system, such as the cellular system described withreference to FIG. 1A and according to the operations previouslydescribed with reference to FIGS. 2–7 and as will subsequently bedescribed with reference to FIGS. 9–10B. The subscriber unit 802includes an RF unit 804, a processor 806 that performs basebandprocessing and other processing operations, and a memory 808. The RFunit 804 couples to an antenna 805 that may be located internal orexternal to the case of the subscriber unit 802. The processor 806 maybe an Application Specific Integrated Circuit (ASIC) or another type ofprocessor that is capable of operating the subscriber unit 802 accordingto the present invention. The memory 808 includes both static anddynamic components, e.g., DRAM, SRAM, ROM, EEPROM, etc. In someembodiments, the memory 808 may be partially or fully contained upon anASIC that also includes the processor 806. A user interface 810 includesa display, a keyboard, a speaker, a microphone, and a data interface,and may include other user interface components. The RF unit 804, theprocessor 806, the memory 808, and the user interface 810 couple via oneor more communication buses/links. A battery 812 also couples to andpowers the RF unit 804, the processor 806, the memory 808, and the userinterface 810.

The RF unit 804 includes the RF transmitter components described withreference to FIG. 2 and operates according to the present invention toadjust the envelope of an RF signal produced by a translational loopcontained therein. The structure of the subscriber unit 802 illustratedis only one particular example of a subscriber unit structure. Manyother varied subscriber unit structures could be operated according tothe teachings of the present invention. Further, the principles of thepresent invention may be applied to base stations, as are generallydescribed with reference to FIG. 1A.

FIG. 9 is a logic diagram illustrating a method of operation accordingto the present invention. Operation according to the present inventionis initiated upon power up or reset. In such case, the operations ofFIG. 9 will be performed along with a number of other operationsrequired to set/reset the operation of a corresponding RF transmitter200. The operations of FIG. 9 will be described with additionalreference to FIGS. 10A and 10B. FIG. 10A is a graph illustrating thepower spectral density of an RF signal generated by the RF transmitter200 of the present invention with the time delay control signalimproperly set. FIG. 10B is a graph illustrating the power spectraldensity of an RF signal generated by the RF transmitter 200 of thepresent invention with the time delay control signal properly set.

Referring again to FIG. 9, upon power up or reset, the calibrationoperations of the present invention are initiated (step 902). In suchcase, the components of the RF transmitter 200, e.g., RF transmitter 200of FIG. 2, operate to produce the RF signal. The RF signal may have apower spectral density such as that shown in FIG. 10A. With theoperations of the present invention, the CP (central lobe of the PSD)and the ACP (side lobe of the PSD) are measured (step 904). Then, theratio of the CP to the ACP (or the ACP to the CP) is determined, i.e.,ACPR. The ACPR is then compared to an ACPR threshold (step 906). ThisACPR threshold relates directly to the permissible interference thatoperation on the channel may produce in the alternate channel (oradjacent channel).

With the PSD shown in FIG. 10A, the ACPR does not meet the threshold andthe time delay control signal provided by the time delay calibrationblock 214 to the envelope time delay adjust block 212 is adjusted (step908). Operation returns from step 908 to step 904 where the CP and ACPare again measured and the ACPR is determined. After one or moreiterations of step 908, the PSD of FIG. 10B is produced such that theACPR threshold is met at step 906. From step 906, with the ACPR met,operation proceeds to step 910 wherein the currently set time delaycontrol signal produced by the time delay calibration block 214 is used.Operation remains at step 912 until a calibration event occurs. Acalibration event may occur periodically or when a threshold is met.

FIG. 11 is a graph illustrating the phase response of a translationalloop employed by an RF transmitter constructed according to the presentinvention. As shown in FIG. 11, the translational loop, e.g.,translational loop 204 of FIG. 2, has a phase response that varies withfrequency. Within the desired channel, the phase response of thetranslational loop is substantially constant. Further, within anadjacent channel, the phase response of the translational loop isconstant within a portion of the adjacent channel but varying withfrequency in another portion of the adjacent channel. Within analternate channel, the phase response of the translational loop variessubstantially with frequency. Other translational loops may exhibitdifferent phase response that varies greater with frequency or lesserwith frequency. The graph of FIG. 11 is shown only to describe theteachings of the present invention.

In order to reduce/minimize the adjacent channel power and alternatechannel power produced by the translational loop, the phase and envelopeof the translational loop must be closely matched for all frequencies,including the desired channel, the adjacent channel, and the alternatechannel. The teachings of the present invention, as described withreference to FIGS. 2–10 show how this goal may be fully accomplished fortranslational loops having a constant phase response and mostlyaccomplished for translational loops having a phase response that varieswith frequency. However, for translational loops exhibiting the behaviorof FIG. 11, the phase and envelope of the translational loop cannot befully matched over the full frequency spectrum of the desired channel,the adjacent channel, the alternate channel. Because envelope and phasematching can only be fully performed within the desired channel (havinga constant phase response), adjacent channel power and alternate channelpower cannot be fully minimized for translational loops exhibiting thephase response of FIG. 11.

Thus, according to the present invention, adjacent channel and alternatechannel components of the baseband signal are phase equalized prior totheir application to the translational loop. In one particular operationaccording to the present invention, a baseband signal is phase equalizedaccording to the phase equalization curve shown in FIG. 11 prior to itsconversion to IF. Phase equalization of the baseband signalpre-conditions the baseband signal to overcome the operationalshortcomings introduced by the translational loop due to its phaseresponse. In effect, phase equalizing the baseband signal counteractsthe phase response of the translational loop. Thus, when the envelopeand phase of the output of the translational loop are matched, adjacentchannel power and alternate channel power are significantly reduced.

FIG. 12 is a block diagram illustrating another embodiment of an RFtransmitter constructed according to the present invention. Thestructure of the RF transmitter 1200 is similar to the structure of theRF transmitter 200 described with reference to FIG. 2 and elementscommon to FIG. 2 retain common numbering. However, the RF transmitter1200 of FIG. 12 further includes a phase equalizer 1202 and a Digital toAnalog Converter (DAC) 1204. The phase equalizer 1202 receives amodulated baseband signal from the baseband processor and produces aphase equalized modulated baseband signal. Phase equalization performedby the phase equalizer 1202 is performed so as to produce a phaseequalization curve illustrated in FIG. 11, for example. The actual phaseequalization that the phase equalizer 1202 produces is based uponcalibration operations performed on the translational loop 204. Thesecalibration operations will be described further with reference to FIG.15.

With this structure of the RF transmitter 1200, the phase equalizer 1202receives digital I and Q components of the modulated baseband signalfrom a coupled baseband processor. The phase equalizer 1202 phaseequalizes the digital modulated baseband signal and produces a digitalphase equalized modulated baseband signal (including both I and Qcomponents). The DAC 1204 receives the digital phase equalized modulatedbaseband signal and converts the digital phase equalized modulatedbaseband signal to an analog phase equalized modulated baseband signal(including both I and Q components) that it provides to the IF modulator202.

In this digital embodiment of the phase equalizer 1202, a digital allpass filter that operates in the time domain may be employed. Such astructure is described further with reference to FIG. 14. Alternately,the operations of the phase equalizer 1202 may be performed in thefrequency domain. A structure that performs phase equalization in thefrequency domain is described with reference to FIG. 13.

In an alternate construction of the RF transmitter, the phase equalizer1202 is an analog component that performs analog phase equalizationoperations. In such case, the phase equalizer 1202 would receive ananalog modulated baseband signal from the coupled baseband processor andproduce an analog phase equalized modulated baseband signal. In oneconstruction in this embodiment, the phase equalizer 1202 is an analogall pass filter having a phase response that corresponds to the phaseequalization curve of FIG. 11.

With the RF transmitter 1200 of FIG. 12, a DAC 1206 is shown to beexternal to the envelope time delay adjust block 212. With thisstructure, the envelope time delay adjust block 212 produces a digitaltime delayed envelope signal that is converted to an analog equivalentthereof by the DAC 1206. The output of the DAC 1206 is then low passfiltered by low pass filter 1208 prior to its application to envelopeadjust block 206.

As was the case with the RF transmitter 200 of FIG. 2, with the RFtransmitter 1200 of FIG. 12, the envelope detection block 202 mayreceive input from three separate sources. In a first operation, theenvelope detection block 202 receives its input from the basebandprocessor. In a second operation, the envelope detection block 202receives its input from the output of the phase equalizer 1202. In athird operation, the envelope detection block 202 receives its inputfrom the output of the IF modulator 202.

FIG. 13 is a block diagram illustrating one embodiment of a phaseequalizer of the RF transmitter of FIG. 12 constructed according to thepresent invention. With the phase equalizer 1300 of FIG. 13, phaseequalization is done in the frequency domain upon digital signals. Thephase equalizer 1300 includes Fast Fourier Transform (FFT) operationalblock 1302, phase adjust block 1304, and Inverse Fast Fourier Transform(IFFT) operational block 1306 that operate upon the I component of thedigital modulated baseband signal. Likewise, the phase equalizer 1300includes Fast Fourier Transform (FFT) operational block 1308, phaseadjust block 1310, and Inverse Fast Fourier Transform (IFFT) operationalblock 1312 that operate upon the Q component of the digital modulatedbaseband signal.

The FFT operational blocks 1302 and 1304 receive the I and Q componentsof the digital modulated baseband signal in the time domain,respectively, and convert the I and Q components of the digitalmodulated baseband signal to the frequency domain. The phase adjustblock 1304 and the phase adjust block 1310 receive phase equalizercalibration settings 1314 and, based upon the phase equalizercalibration settings 1314, adjust the I and Q components of themodulated baseband signal in the frequency domain to produce I and Qcomponents of the phase equalized modulated baseband signal in thefrequency domain. Then, IFFT operational blocks 1306 and 1312 receivethe I and Q components of the phase equalized modulated baseband signalin the frequency domain and produce the phase equalized modulatedbaseband signal in the time domain. The phase equalizer calibrationsettings 1314 cause the phase adjust blocks 1304 and 1310 to implementthe phase equalization curve illustrated in FIG. 11.

FIG. 14 is a block diagram illustrating another embodiment of a phaseequalizer of the RF transmitter of FIG. 12 constructed according to thepresent invention. The phase equalizer includes all pass filter 1402,all pass filter 1404, and phase equalizer calibration settings 1406. Allpass filter 1402 receives the I component of the modulated basebandsignal and produces the I component of the phase equalized modulatedsignal. All pass filter 1404 receives the Q component of the modulatedbaseband signal and produces the Q component of the phase equalizedmodulated signal. All pass filter 1402 and all pass filter 1404 receiverespective phase equalizer calibration settings and perform phaseequalization operations based thereon. All pass filters are well knownin the art. For one description of all pass filters, see Adel S. Sedraand Kenneth C. Smith, Microelectronic Circuits, Fourth Edition, OxfordUniversity Press, 1998.

FIG. 15 is a logic diagram illustrating a method for calibrating a phaseequalizer according to the present invention. Calibration may beinitiated at power up, reset, periodically, or otherwise (step 1502).Because the operational characteristics of the components of the RFtransmitter including the translational loop may change withtemperature, periodic calibration may be preferred. With calibrationoperations commenced, a first selected desired channel is chosen (step1504). Then, with the RF transmitter transmitting in the desired channelwith a test signal, as provided by the baseband processor, the phaseresponse of the translational loop within the adjacent and alternatechannels are measured (step 1504).

If the chosen desired channel is the last desired channel to beconsidered, operation proceeds to step 1510. If not, operation returnsto step 1504 wherein the next desired channel is selected. Typically,the phase response of the translational loop varies not only with thedifference in frequency between the desired channel andadjacent/alternate channels but also with the frequency of the desiredchannel. As such, calibration operations are performed for each desiredchannel that is serviced by the translational loop. However, with sometranslational loops, the phase response may be only a function of thedifference between the desired channel and the adjacent/alternatechannels. In such case, only a single desired channel is considered inthe operations of FIG. 15.

With measurements obtained for each desired channel, operation proceedsto step 1510 in which phase equalizer calibration settings aredetermined. The phase equalizer calibration settings determined will bebased upon the structure of the phase equalizer employed. When the phaseequalizer is an all pass filter, the phase equalizer calibrationsettings will include filter coefficients. When the phase equalizeremploys frequency domain operations, the phase equalizer calibrationsettings may include complex frequency domain coefficients.

FIG. 16 is a logic diagram illustrating operation of a phase equalizeraccording to the present invention. RF transmissions at any time will bewithin a desired channel. Thus, as a first operation, the desiredchannel is selected and the local oscillator is set accordingly (step1602). Then, the phase equalizer settings for the desired channel areselected and retrieved (step 1604). The phase equalizer settings arethen loaded into the phase equalizer (step 1606). Operation continueswith these phase equalizer settings until the desired channel changes(step 1608) at which point, a new desired channel is selected (step1610) and operation proceeds to step 1602. In another embodiment, asingle set of equalizer calibration settings are employed for alldesired channels.

The invention disclosed herein is susceptible to various modificationsand alternative forms. Specific embodiments therefore have been shown byway of example in the drawings and detailed description. It should beunderstood, however, that the drawings and detailed description theretoare not intended to limit the invention to the particular formdisclosed, but on the contrary, the invention is to cover allmodifications, equivalents and alternatives falling within the spiritand scope of the present invention as defined by the claims.

1. A Radio Frequency (RF) transmitter comprising: a phase equalizer thatreceives a modulated baseband signal and that produces a phase equalizedmodulated baseband signal; an Intermediate Frequency (IF) modulator thatreceives the phase equalized modulated baseband signal and that producesa modulated IF signal having a non-constant envelope; a translationalloop that receives the modulated IF signal and that produces a modulatedRF signal having a constant envelope; an envelope time delay adjustblock that receives an envelope signal corresponding to the modulated IFsignal and that produces a time delayed envelope signal based upon atime delay control signal; an envelope adjust block that adjusts themodulated RF signal based upon the time delayed envelope signal toproduce an envelope adjusted modulated RF signal; and a time delaycalibration block that receives the envelope adjusted modulated RFsignal and that produces the time delay control signal.
 2. The RFtransmitter of claim 1, wherein the time delay calibration blockcomprises: a down converter that converts the envelope adjustedmodulated RF signal to a baseband signal; an Analog to Digital Converter(ADC) that samples the baseband signal; a Low Pass Filter (LPF) thatfilters the baseband signal to produce a LPF output; a Band Pass Filter(BPF) that filters the baseband signal to produce a BPF output; and alevel detector and control block that receives the LPF output and theBPF output and that produces the time delay control signal based uponthe LPF output and the BPF output.
 3. The RF transmitter of claim 2,wherein the BPF comprises: a complex mixer; and a LPF.
 4. The RFtransmitter of claim 1, wherein the time delay calibration block:determines a channel power corresponding to the RF signal; determines analternate channel power corresponding to an alternate channel or anadjacent channel; and determines the time delay control signal basedupon a ratio of the channel power and the alternate channel power oradjacent channel power.
 5. The RF transmitter of claim 1, furthercomprising an envelope detection block that produces the envelopesignal.
 6. The RF transmitter of claim 5, wherein the envelope detectionblock determines the envelope signal based upon the phase equalizedmodulated baseband signal.
 7. The RF transmitter of claim 5, wherein theenvelope detection block determines the envelope signal based upon themodulated IF signal.
 8. The RF transmitter of claim 5, wherein theenvelope detection block receives the envelope signal from a coupledbaseband processor.
 9. The RF transmitter of claim 1, wherein: theenvelope signal is a digital signal; and the time delayed envelopesignal is an analog signal.
 10. The RF transmitter of claim 9, whereinthe envelope time delay adjust block comprises: a time delay block thatdelays the digital envelope signal by a delay that is based upon thetime delay control signal; and a digital to analog converter thatreceives the output of the time delay block and that produces the timedelayed envelope signal.
 11. The RF transmitter of claim 1: wherein themodulated baseband signal is a digital signal; and wherein the phaseequalized modulated baseband signal is an analog signal.
 12. The RFtransmitter of claim 1: wherein the modulated baseband signal is adigital signal; wherein the phase equalized modulated baseband signal isan digital signal; and the RF transmitter further comprises a digital toanalog converter (DAC) coupled between the phase equalizer and the IFmodulator, wherein the DAC receives the digital phase equalizedmodulated baseband signal from the phase equalizer, converts the digitalphase equalized modulated baseband signal to an analog phase equalizedmodulated baseband signal, and provides the analog phase equalizedmodulated baseband signal to the IF modulator.
 13. The RF transmitter ofclaim 1, wherein the phase equalizer comprises an all-pass filter thatadjusts the phase of the modulated baseband signal within at least analternate channel.
 14. The RF transmitter of claim 13, wherein theall-pass filter is a digital filter.
 15. The RF transmitter of claim 13,wherein the all-pass filter is an analog filter.
 16. The RF transmitterof claim 1, wherein the phase equalizer comprises: a Fast FourierTransform (FFT) operation; a phase adjust block; and an Inverse FastFourier Transform (IFFT) operation.
 17. The RF transmitter of claim 16,wherein the phase adjust block adjusts the phase of the modulatedbaseband signal within at least an alternate channel.
 18. The RFtransmitter of claim 16, wherein the phase adjust block adjusts thephase of the modulated baseband signal within at least an adjacentchannel.
 19. A wireless device comprising: a case; an antenna coupled tothe case; a baseband processor disposed within the case; a RadioFrequency (RF) unit disposed within the case, coupled to the basebandprocessor, coupled to the antenna and having an RF transmittercomprising: a phase equalizer that receives a modulated baseband signalfrom the baseband processor and that produces a phase equalizedmodulated baseband signal; an Intermediate Frequency (IF) modulator thatreceives the phase equalized modulated baseband signal and that producesa modulated IF signal having a non-constant envelope; a translationalloop that receives the modulated IF signal and that produces a modulatedRF signal having a constant envelope; an envelope time delay adjustblock that receives an envelope signal corresponding to the modulated IFsignal and that produces a time delayed envelope signal based upon atime delay control signal; an envelope adjust block that adjusts themodulated RF signal based upon the time delayed envelope signal toproduce an envelope adjusted modulated RF signal; and a time delaycalibration block that receives the envelope adjusted modulated RFsignal and that produces the time delay control signal.
 20. The wirelessdevice of claim 19, wherein the time delay calibration block comprises:a down converter that converts the envelope adjusted modulated RF signalto a baseband signal; an Analog to Digital Converter (ADC) that samplesthe baseband signal; a Low Pass Filter (LPF) that filters the basebandsignal to produce a LPF output; a Band Pass Filter (BPF) that filtersthe baseband signal to produce a BPF output; and a level detector andcontrol block that receives the LPF output and the BPF output and thatproduces the time delay control signal based upon the LPF output and theBPF output.
 21. The wireless device of claim 19, wherein the time delaycalibration block: determines a channel power corresponding to the RFsignal; determines an alternate channel power corresponding to analternate channel or an adjacent channel; and determines the time delaycontrol signal based upon a ratio of the channel power and the alternatechannel power.
 22. The wireless device of claim 19, further comprisingan envelope detection block that produces the envelope signal.
 23. Thewireless device of claim 22, wherein the envelope detection blockdetermines the envelope signal based upon the phase equalized modulatedbaseband signal.
 24. The wireless device of claim 22, wherein theenvelope detection block determines the envelope signal based upon themodulated IF signal.
 25. The wireless device of claim 22, wherein theenvelope detection block receives the envelope signal from a coupledbaseband processor.
 26. The wireless device of claim 19, wherein: theenvelope signal is a digital signal; and the time delayed envelopesignal is an analog signal.
 27. The wireless device of claim 26, whereinthe envelope time delay adjust block comprises: a time delay block thatdelays the digital envelope signal by a delay that is based upon thetime delay control signal; and a digital to analog converter thatreceives the output of the time delay block and that produces the timedelayed envelope signal.
 28. The wireless device of claim 19: whereinthe modulated baseband signal is a digital signal; and wherein the phaseequalized modulated baseband signal is an analog signal.
 29. Thewireless device of claim 19: wherein the modulated baseband signal is adigital signal; wherein the phase equalized modulated baseband signal isan digital signal; and the RF transmitter further comprises a digital toanalog converter (DAC) coupled between the phase equalizer and the IFmodulator, wherein the DAC receives the digital phase equalizedmodulated baseband signal from the phase equalizer, converts the digitalphase equalized modulated baseband signal to an analog phase equalizedmodulated baseband signal, and provides the analog phase equalizedmodulated baseband signal to the IF modulator.
 30. The wireless deviceof claim 19, wherein the phase equalizer comprises an all-pass filterthat adjusts the phase of the modulated baseband signal within at leastan alternate channel.
 31. The wireless device of claim 30, wherein theall-pass filter is a digital filter.
 32. The wireless device of claim30, wherein the all-pass filter is an analog filter.
 33. The wirelessdevice of claim 19, wherein the phase equalizer comprises an all-passfilter that adjusts the phase of the modulated baseband signal within atleast an adjacent channel.
 34. The wireless device of claim 19, whereinthe phase equalizer comprises: a Fast Fourier Transform (FFT) operation;a phase adjust block; and an Inverse Fast Fourier Transform (IFFT)operation.
 35. The wireless device of claim 34, wherein the phase adjustblock adjusts the phase of the modulated baseband signal within at leastan alternate channel.
 36. The wireless device of claim 34, wherein thephase adjust block adjusts the phase of the modulated baseband signalwithin at least an adjacent channel.
 37. A method for producing amodulated RF signal having a non-constant envelope, the methodcomprising: receiving a modulated baseband signal; phase adjusting themodulated baseband signal to produce a phase equalized modulatedbaseband signal; converting the phase equalized modulated basebandsignal to a modulated IF signal having a non-constant envelope;converting the modulated IF signal to a modulated RF signal having aconstant envelope using a translational loop; receiving an envelopesignal corresponding to the modulated IF signal; producing a timedelayed envelope signal based upon a time delay control signal;adjusting the modulated RF signal based upon the time delayed envelopesignal to produce an envelope adjusted modulated RF signal that has anon-constant envelope; and producing the time delay control signal basedupon the envelope adjusted modulated RF signal.
 38. The method of claim37, wherein producing the time delay control signal based upon theenvelope adjusted modulated RF signal comprises: converting the envelopeadjusted modulated RF signal to an envelope adjusted modulated basebandsignal; low pass filtering the envelope adjusted modulated basebandsignal to produce a low pass filtered output; band pass filtering theenvelope adjusted modulated baseband signal to produce a band passfiltered output; and determining the time delay control signal basedupon the ratio of the band pass filtered output to the low pass filteredoutput.
 39. The method of claim 37, wherein producing the time delaycontrol signal based upon the envelope adjusted modulated RF signalcomprises: determining a channel power corresponding to the RF signal;determining an alternate channel power corresponding to an alternatechannel or an adjacent channel; and determining the time delay controlsignal based upon a ratio of the channel power and the alternate channelpower.
 40. The method of claim 37, further comprising determining theenvelope signal based upon the phase equalized baseband signal.
 41. Themethod of claim 37, further comprising determining the envelope signalbased upon the modulated IF signal.
 42. The method of claim 37, furthercomprising receiving the envelope signal from a coupled basebandprocessor.
 43. The method of claim 37: wherein the modulated basebandsignal is a digital signal; and wherein the phase equalized modulatedbaseband signal is an analog signal.
 44. The method of claim 37: whereinthe modulated baseband signal is a digital signal; wherein the phaseequalized modulated baseband signal is an digital signal; and whereinphase adjusting the modulated baseband signal to produce a phaseequalized modulated baseband signal is performed digitally.
 45. Themethod of claim 37, wherein phase adjusting the modulated basebandsignal to produce a phase equalized modulated baseband signal isperformed using an all-pass filter.
 46. The method of claim 37, whereinphase adjusting the modulated baseband signal to produce a phaseequalized modulated baseband signal is performed by: converting themodulated baseband signal from the time domain to the frequency domain;phase adjusting the modulated baseband signal to produce the phaseequalized modulated baseband signal in the frequency domain; andconverting the phase equalized modulated baseband signal from the timedomain to the frequency domain.
 47. The method of claim 37, whereinphase adjusting the modulated baseband signal to produce the phaseequalized modulated baseband signal includes adjusting the phase of themodulated baseband signal within at least an alternate channel.
 48. Themethod of claim 37, wherein phase adjusting the modulated basebandsignal to produce the phase equalized modulated baseband signal includesadjusting the phase of the modulated baseband signal within at least anadjacent channel.